Transmitter identification system

ABSTRACT

The invention relates to a transmitter identification system, which utilizes an identification signal embedded into a digital television signal, enabling the transmitter of origin to be identified at a receiving station. Ideally the identification signal is an orthogonal pseudo-random sequence time synchronized to the signal frame structure of the digital television signal. Particularly designed for single frequency networks, identification of the various transmitted signals enables the network to be tuned to eliminate or minimize multi-path effects at certain locations, which receive transmissions from various transmitters.

CROSS-REFERENCE TO RELATED APPLICATIONS

The present application is a continuation-in-part of U.S. patent application Ser. No. 10/765,138 filed Jan. 28, 2004, which claims priority from U.S. patent application Ser. No., 60/443,550 filed Jan. 30, 2003, which are all incorporated herein by reference.

TECHNICAL FIELD

The present invention relates to a transmitter identification system, and in particular to a digital television (DTV) transmitter identification system for identifying the origin of a received DTV signal, which can be used for tuning a distributed-transmission (single-frequency) DTV network, geographic locating, estimating the channel impulse response for a particular transmitter with a very long delay spread capacity, and transmitting robust low bit rate control information to mobile and stationary terminals.

BACKGROUND OF THE INVENTION

Digital television (DTV) networks are comprised of a plurality of transmitters, each broadcasting the same signal using multiple frequencies or a single frequency (single frequency network). As the number of transmitters grows, there is an increased desire to be able to identify the transmitter of origin for each signal received. Transmitter identification will enable broadcasting authorities to identify illegal or improperly operating transmitters. Moreover, transmitter identification can also be used to tune various transmitters in a single frequency network to minimize the effects of multi-path interference. Multi-path interference is caused by the destructive interference of several different transmissions originating from different transmitters and/or caused by the reflection of transmissions. FIG. 1 illustrates a single-frequency digital-television network 1, including three transmitters 2, 3 and 4 with transmission ranges 6, 7 and 8, respectively. With reference to the overlap area, designated by reference numeral 9, a receiver positioned therein would receive a transmitted signal comprised of direct transmitted signals 12, 13 and 14 from transmitters 2, 3 and 4, respectively, plus reflected transmitted signal 16 from transmitter 2. The reflected transmitted signal 16 resulting from a reflection off of a large obstacle 17, e.g. a large building. Unfortunately, the various components of the transmitted signal may not all be in phase, resulting in undesired effects. The effects of multi-path interference to DTV signals include a degradation in the television picture and sound. In analog television, multi-path interference causes “ghost” images.

U.S. Pat. Nos. 6,075,823 issued Jun. 13, 2000 to Hideaki Sonoda; 6,122,015 issued Sep. 19, 2000 to Al-Dhahir et al; 6,128,337 issued Oct. 3, 2000 to Schipper et al; 6,304,299 issued Oct. 16, 2001 to Frey et al; 6,437,832 issued Aug. 20, 2002 to Orabb et al; and 6,501,804 issued Dec. 31, 2000 issued to Rudolph et al disclose various solutions to overcoming the problem of multi-path interference. In general, the systems disclosed in the aforementioned references compare a transmitted test signal including noise with a reference signal, and construct a filter in accordance with the results of the comparison to remove noise from transmitted digital television signals. Unfortunately, none of the prior art references provide an identification signal for each transmitter, nor do they provide a system for tuning the entire network. Each of the aforementioned systems requires a complicated filtering circuit to be installed in every receiver in the system, which greatly increase the cost to the operator, and therefore the consumer.

Conventional Global Positioning Systems (GPS) do not work well inside buildings, due to the weak field strength and the high frequency of the GPS signal. In contrast to the GPS signals, the DTV signals are received from transmitters at relatively short distances, and the broadcast transmitters operate at levels up to megawatts of effective radiated power (ERP). Since the locations of DTV transmitters in DTV networks are known, it is possible to locate the position of a receiver, when the DTV signals from multiple DTV transmitters are successfully received and identified. Moreover, the RF frequency of the DTV signal is much lower than that of the GPS, which makes it easier for the DTV signal to penetrate buildings and other objects. Accordingly, there is often sufficient field strength to permit DTV signal reception and position location even inside buildings. Furthermore, the wide bandwidth of the DTV signal help to improve the accuracy of the position location.

An object of the present invention is to overcome the shortcomings of the prior art by providing a transmitter identification system that can be used to identify the transmissions, direct or redirected, from various transmitters.

Another object of the present invention is to provide timing information relating to the transmissions from known transmitters, which can be used to tune the transmitters in a network to minimize the effects of multi-path interference.

Another object of the present invention is to provide receiver location information, based on the relative position of the receiver to a plurality of transmitters.

Another object of the present invention is to provide a robust data transmission system, with a relatively large coverage area, based on the modulation of the transmitter identification sequences.

SUMMARY OF THE INVENTION

Accordingly, the present invention relates to a method of identifying a transmitter in a distributed digital television transmission network, including a plurality of transmitters and a plurality of receivers, comprising the steps of:

a) providing a signal to be transmitted to each transmitter;

b) embedding an identification sequence into the signal, indicative of the transmitter of origin, forming a combined transmission; and

c) transmitting the combined transmission from each transmitter.

BRIEF DESCRIPTION OF THE DRAWINGS

The invention will be described in greater detail with reference to the accompanying drawings which represent preferred embodiments thereof, wherein:

FIG. 1 is a schematic illustration of a conventional Digital Television Network;

FIG. 2 illustrates a DTV signal frame structure including an identification sequence x_(i) synchronized therein;

FIG. 3 illustrates a 16-bit Kasami sequence generator;

FIG. 4 illustrates a ATSC signal data field;

FIG. 5 illustrates an auto-correlation function of a 16-bit Kasami sequence;

FIG. 6 a illustrates a cross-correlation function over a single segment;

FIG. 6 b illustrates a cross-correlation function averaged over 60 segments;

FIG. 7 illustrates an impulse in a cross-correlation function;

FIG. 8 illustrates an impulse in the cross-correlation function after side-lobe filtering; and

FIG. 9 is a schematic illustration of a DTV network including a receiver location determining system.

DETAILED DESCRIPTION

In accordance with the present invention, the transmitter identification system embeds an identification sequence in the form of a pseudo-random sequence x_(i)(n), selected from a set of orthogonal sequences, in band into each DTV signal d_(i)(n) creating a combined transmission d_(i)′(n). In practice, the sequences will be truncated and, therefore, not be perfectly orthogonal; however, for the purposes of the invention they will only need to have negligible cross correlation. Accordingly, orthogonal, substantially-orthogonal and having negligible cross correlation will be used interchangeably so as not to limit the scope of protection to perfectly orthogonal.

The process is represented by the equation: d _(i)′(n)=d _(i)(n)+μx_(i)(n)  (1)

wherein ρ represents a gain coefficient controlling the embedding level of the identification sequence, which varies from transmitter to transmitter depending on the modulation and channel coding schemes of the individual transmitters. After passing through a transmission channel h_(i), a transmitted signal r_(i) from the i^(th) transmitter can be formulated as: r _(i)(n)=d _(i)′(n)⊕h _(i) +n _(i)(n)  (2)

where n_(i)(n) is the noise for the i-th transmitter.

The overall transmitted signal r(n) can be formulated as: $\begin{matrix} {{r(n)} = {\sum\limits_{i = 1}^{M}\left\lbrack {{{d_{i}^{\prime}(n)} \otimes h_{i}} + {n_{i}(n)}} \right\rbrack}} & (3) \end{matrix}$

Identification of a particular transmitter is impossible without additional identification processes. According to the present invention, details of the existence of a specific transmitter and the strength of each transmitted signal r_(i)(n) at the reception site can be determined by calculating correlating functions. For example, the correlation between r(n) and a locally generated identification signal x_(j)(n) can provide identifying information, i.e. existence and strength of the signal, about the j-th transmitter. If a signal from the j-th transmitter is present, i.e. the transmitted signal r(n) contains the identification sequence x_(i)(n) matching the locally generated sequence x_(j)(n), an impulse will appear in the c ross correlation function (see FIG. 5). If more than one impulse is found for a given sequence, the impulse separations are indicative of a multi-path delay. Accordingly, this method can be used in obtaining the impulse response from each particular transmitter.

For a single frequency network, in which each transmitter transmits at the same frequency, the signal from each transmitter can be independently tuned, e.g. for power level and relative time delay between transmitters, so that the effects of multi-path interference are minimized in overlap areas, see area 9 in FIG. 1. At a given test station inside the overlap area, the cross-correlation functions for the various transmitters are compared, and the relative power levels of the signals from each transmitter are determined. From this information, it is possible to minimize multi-path effects by either delaying the transmission from one or more of the transmitters relative to one or more of the others, thereby maximizing the number of the signals that are received synchronously, or by adjusting the power level of one or more transmitter signals to lesson or increase their effect on the overall transmitted signal. The tuning will have minimal or no effect on the reception at various locations outside the overlap areas, but can greatly improve the reception at locations inside the overlap areas. Preferably, the comparisons are carried out at a plurality of test stations within the overlap area, and the transmitters are tuned in such a manner as to optimize the reception throughout the overlap area.

The cross correlation between r(n) and x_(j)(n) is defied by $\begin{matrix} \begin{matrix} {{R_{{rx}_{j}}(m)} = {\sum\limits_{n = 0}^{N - 1}{{r(n)}{x_{j}\left( {n - m} \right)}}}} \\ {= {\sum\limits_{n = 0}^{N - 1}{\left\{ {{\sum\limits_{i = 1}^{M}{{d_{i}^{\prime}(n)} \otimes h_{i}}} + {n_{i}(n)}} \right\}{x_{j}\left( {n - m} \right)}}}} \\ {= {\sum\limits_{n = 0}^{N - 1}{\left\{ {\sum\limits_{i = 1}^{M}\left\lbrack {{\left( {{d_{i}(n)} + {\rho\quad{x_{i}(n)}}} \right) \otimes h_{i}} + {n_{i}(n)}} \right\rbrack} \right\}{x_{j}\left( {n - m} \right)}}}} \\ {= {{\rho\quad{R_{x_{j}x_{j}} \otimes h_{j}}} + {\sum\limits_{{i = 1},{i \neq j}}^{M}{\rho\quad{R_{x_{i}x_{j}} \otimes h_{i}}}} +}} \\ {\sum\limits_{n = 0}^{N - 1}{\sum\limits_{i = 1}^{M}{\left\lbrack {{d_{i}(n)} + {n_{i}(n)}} \right\rbrack{x_{j}\left( {n - m} \right)}}}} \end{matrix} & (4) \end{matrix}$

With the orthogonal property of the selected sequence, the autocorrelation functions R_(xjxj) of the locally generated identification sequence x_(j), can be approximated as a delta function. The second and third terms in the above equation (4) are only noise like sequences from the in-band DTV signals of the same transmitter and other transmitters. Therefore, the received channel response h_(j) from the j-th transmitter can be approximated by R_(rxj), i.e. R _(rxj)(m)=Ah _(j)+noise  (5)

where A is a constant determined by R_(xjxj) and the gain coefficient ρ. The received channel response h_(j) from the j-th transmitter can be determined as R_(xjxj) and ρ are known.

The earliest correlation peak that exceeds a particular threshold is selected as corresponding to the most direct line of sight signal path. The position of this earliest correlation peak can then be converted to relative propagation time in terms of seconds, as other correlation peaks arrive, to determine the location of the reception point, as will be hereinafter discussed. The correlation functions in (4) and (5) can be interpolated to improve the precision of the propagation time determined.

With reference to FIG. 2, the identification sequence x_(i) is time synchronized to the DTV signal frame structure. The illustrated signal relates specifically to an Advanced Television Systems Committee (ATSC) DTV system, but the invention is applicable to any similar system, e.g. Digital Video Broadcasting-Terrestrial (DVB-T) or Integrated Services Digital Broadcasting-Terrestrial (ISDB-T) systems.

Different injection levels of the embedded identification sequence x_(i) are determined for ATSC, DVB-T and ISDB systems, respectively. For ATSC systems, Kasami sequences are buried between 10 dB to 30 dB below the DTV system noise threshold, which causes negligible impact to DTV signal reception.

Preferably, 16-bit Kasami sequences are used as identification sequences for a North American ATSC DTV system. However, Gold sequences and any other suitable substantially-orthogonal pseudo-random sequences may be used. The use of 16-bit Kasami sequence is a compromise of the sequence length, spreading gain and the number of the sequences, which are available for DTV transmitter identification. FIG. 3 illustrates a 16-bit Kasami sequence generator, in which there are 16+8=24 digits or 2²⁴−1 different initial states, which results in 2²⁴−1 different Kasami sequences. With reference to FIG. 4, each ATSC signal data field has 312×832=259,584 symbols (including segment synchronization), therefore, three complete 16-bit Kasami sequences (2¹⁶−1=65535 chips) and one truncated 16-bit Kasami sequence (2¹⁶−1−2519=63016 chips) can be fitted into one ATSC field. The Kasami sequence chip rate should be the same as the ATSC DTV system symbol rate, i.e. 10.7622378 Msps. The Kasami sequences are injected during the transmission of the DTV data segments, but not during the ATSC DTV field synchronization transmission period to avoid interference with DTV signal acquisition.

The transmitter identification process can be further reduced, if the initial values for the sequence generators only differ in the last few bits for the neighboring transmitters. By assigning different Kasami sequences this way, a blind search approach can be avoided during the transmitter identification process.

Since the 16-bit Kasami sequence is very long and takes a long time to synchronize, it would be advantageous if a smaller portion of the DTV signal could be identified as a starting point, thereby facilitating synchronization. In North America the ATSC DTV field sync. PN-511 sequence, which has high signal strength, can be used as a “short code” for quick detection and synchronization of the Kasami sequence. For DVB-T and ISDB-T systems, in Europe and Japan, the cyclic prefix of the OFDM symbol can be used. Furthermore, rather than correlate the entire Kasami sequence with the received signal, the correlation function can be calculated only between the PN-511 sequence (or the cyclic prefix of the DVB-T and ISDB-T signals) and the received signal.

To reduce the computation complexity during the transmitter identification process, only a desired portion of the correlation functions between the transmitted signal r(n) and the local identification sequence x(n) is computed. For the complete computation of the cross-correlation between the transmitted signal r(n) and the local identification sequence x(n), the following equation can be used: $\begin{matrix} {{{R(m)} = {\sum\limits_{n = 0}^{N - 1}\left\lbrack {{r\left( {n_{0} + n + m} \right)} \cdot {x(n)}} \right\rbrack}},{m = 0},1,2,{{\ldots\quad N} - 1}} & (6) \end{matrix}$

where n₀ is the starting point of the received signal for correlation computation. For transmitter identification purposes, R(m) is only needed for a length of the maximum delay spread of all the multi-path delays from all of the transmitters. In the terrestrial DTV distributed transmission case, about a 6000 DTV symbol duration or 558 μs is adequate. In fact, between 40 μs and 600 μs would suffice. This is less than 1% of the total cross-correlation function samples, which significantly reduces the computation time.

Rather than conducting the correlation computation continuously in real time, a segment of the transmitted DVB-T, ISDB-T or ATSC DTV signal r(n) can be separated therefrom, each of which contains one complete embedded sequence, for correlation computation.

Upon synchronization of the embedded and locally generated identification sequences, using a PN511 sequence for ATSC signals or a cyclic prefix for DVB-T and ISDB-T signals, the received DTV signal can be divided into segments, each with a length of a DTV field plus two times the delay spread of the channel impulse response. Each segment begins at the starting point of each DTV field minus one delay spread and ends at the stopping point of the DTV field plus one delay spread. A sliding window technique can then be used to select portions of the transmitted signal for calculating the correlation function. The length of the sliding window is identical to one DTV field. As the window slides over the signal segment, the local identification sequence x_(j)(n) is correlated to the received signal portion, which falls into the sliding window.

Time-domain a veraging is a technique used to reduce the in-band A TSC DTV signal interference. Post processing using ensample averaging over several cross-correlation functions can improve the dynamic range of the cross-correlation function, as in FIGS. 6 a and 6 b. Several segments are correlated and an average is taken to cancel out noise distinctive of each segment and to improve resolution. Averaging improves the capability of the detection of co-channel interference and the dynamic range of the impulse response. To reduce the synchronization error effect and to optimize the superimposition of the correlation functions, prior to averaging, the peaks in each correlation function are aligned in amplitude and phase.

With reference to FIG. 7, due to a 6/7/8 MHz DTV bandwidth limit, each impulse in the cross-correlation function is in the form of a sin(x)/x function rather than a delta function. A first sidelobe, about 17 dB below the main lobe, could be misidentified as a multi-path reflection, especially when close-in echoes exist. Post processing, or filtering using an appropriate filter response, over the cross-correlation function can reduce the side lobe of the s in(x)/x function to a negligible level, see FIG. 8. One possible way to 'resolve the band-limitation problem is to eliminate the shape of the non-ideal auto-correlation function from the preliminary channel estimation results. To simplify the notations, the correlation result is written as: R′=Ah+noise  (7)

Where h is the ideal impulse response to be estimated and R′ is the correlation function with a truncated length L′. R′=[R(1), R(2), . . . R(L′)]^(T)  (8)

Where A is determined from the side lobe matrix $A = \begin{bmatrix} {{R_{ww}(L)},} & {{R_{ww}\left( {L - 1} \right)},\ldots} & {R_{ww}(1)} \\ {{R_{ww}\left( {L + 1} \right)},} & {{R_{ww}(L)},\ldots} & {R_{ww}(2)} \\ {{R_{ww}\left( {L + 2} \right)},} & {{R_{ww}\left( {L + 1} \right)},\ldots} & {R_{ww}(3)} \\ \ldots & \ldots & \ldots \\ {{R_{ww}\left( {L + L^{\prime} - 1} \right)},} & {{R_{ww}\left( {L + L^{\prime} - 2} \right)},\ldots} & {R_{ww}(L)} \end{bmatrix}$

when noise is Gaussian noise, h can be resolved using: h=(A ^(H) A)⁻¹ A ^(H) R′  (9)

By inverting the amplitude of the embedded Kasami sequence, one-bit information can be transmitted per Kasami sequence or several Kasami sequences can be used to represent one bit, depending on the injection level of the Kasami sequence. At the receiver, a positive correlation would indicate a ‘1’ and a negative correlation would indicate a ‘0’. This technique can be used to transmit low speed data over the entire DTV coverage area to provide data service or for cue and control.

A robust data transmission system can be implemented by modulating the amplitude of the ATSC transmitter identification watermark. Due to the vestigial side band (VSB) modulation of ATSC signals, only one-dimensional amplitude modulation can be used for the transmitter identification sequences. The constellation of the proposed modulation scheme is also optimized to minimize the bit error rate and the impact on the transmitter identification process.

The data stream that is transmitted can be split into two streams: one low-speed, robust data stream and another high-speed data stream. The low-speed stream, which can be used for possible mobile reception, is generated using a binary phase shift keying (BPSK) type of polarity modulation over the entire watermark signal injected within each DTV field. At the receiver, a positive correlation would indicate a ‘1’ and a negative correlation would indicated a ‘0’. The high-speed data stream can be used for fixed reception and is generated using amplitude modulation over each of the four identical sequences that make up a single watermark in a DTV field. As hereinbefore discussed, each transmitter ID sequence within a frame is made up of four identical kasami sequences, the last sequence being truncated to fit the DTV frame. Correlation impulse amplitudes at the receiver are varied by varying the injection level of the four kasami sequences. For an ATSC system, it is possible to inject a watermark at two different levels plus polarity. In this way, more bit information can be transmitted, i.e. four levels of correlation peaks enables 2-bit data to be transmitted by one Kasami sequence. The lower speed transmission mode can be regarded as a special case of the high speed transmission mode with a lower order modulation scheme To protect the transmitted information from any possible errors, several fields of the DTV signal are combined together to use a linear block code or convolutional code with error correction capability.

The major advantages of the low-speed data transmission system are robustness and an extremely large coverage. The data stream can be used to carry timing offset information, i.e. the difference in start time of the various transmitters in the network, as discussed hereinafter, and/or control information for the distributed transmitters. Alternatively, other information like traffic control, weather information, or public bulletin board data in a large metropolitan area can be transmitted.

Synchronous transmission of the DTV signal and the Kasami sequences is not mandatory for transmitter identification; however, synchronization between the DTV signal d(n) and the embedded transmitter identification code x(n) can reduce the amount of the computation during the identification process significantly, as the PN-511 sequence in the d(n) can provide an accurate starting point of x(n) using some auto-correlation techniques. To achieve the synchronized transmission between the DTV signal and the embedded TxID sequence, a Kasami sequence of M=2¹⁶−1 has to be truncated by 639 symbols such that four truncated codes can be embedded into DTV field. Then each TxID sequences can be modulated with the incoming data.

Assuming: a) the data stream to be transmitted is in polarity form +1 and −1 instead of 1 and 0; b) the DTV signals for the i-th transmitter before and after the injection of the 16-bit Kasami sequence y_(i)(n) are d_(i)(n) and d_(i)′(n), respectively, (Please note y_(i)(n) contains three identical TxID sequence with length of M=2¹⁶−1 followed by one truncated TxID sequence with length of M′=2¹⁶−1−639×4); and c) n is the index for discrete signals, the injected process is: $\begin{matrix} \begin{matrix} {{d_{i}^{\prime}(n)} = {{d_{i}(n)} + {\rho_{l}\quad{\sum\limits_{l = 1}^{4}{D_{i,l}{\Pi_{l}(n)}{y_{i}(n)}}}}}} \\ {where} \\ {{\Pi_{l}(n)} = \left\{ {\begin{matrix} {1,} & {1 \leq {n - {\left( {l - 1} \right)\left( {2^{16} - 1} \right)}} \leq {l\left( {2^{16} - 1} \right)}} \\ {0,} & {else} \end{matrix},{l = 1},2,3,} \right.} \\ {and} \\ \begin{matrix} {{\Pi_{l}(n)} = \left\{ \begin{matrix} {1,} & {{{{3 \times \left( {2^{16} - 1} \right)} + 1} \leq n \leq {832 \times 312}},} \\ {0,} & {else} \end{matrix} \right.} & {l = 4} \end{matrix} \end{matrix} & (10) \end{matrix}$

D_(i,j) is the polarity data to be transmitted, and ρ₁ is the gain coefficient to control the injection gain of the l-th identification sequences within the same DTV field, which can be different since the last TxID sequence is truncated. To achieve the equal correlation peaks at the receiver side, ρ₄ is larger than the other three coefficients ρ₁ to ρ₃. After passing through the channel h_(i), the received signal from the i-th transmitter r_(i) can be formulated as: r _(i)(n)=d _(i)′(n)⊕h _(i) +w _(i)(n)  (11)

where w_(i)(n) is the noise for the i-th transmitter during the propagation. The overall received signal r(n) can be formulated as: $\begin{matrix} {{r(n)} = {\sum\limits_{i = 1}^{T}\left\lbrack {{{d_{i}^{\prime}(n)} \otimes h_{i}} + {w_{i}(n)}} \right\rbrack}} & (12) \end{matrix}$

where T is the total number of the transmitters.

To demodulate the polarity data D_(i,l), the received signal r for each field first has to be divided into four r_(l) (l=1, 2, 3, 4), where r _(l)=ℑ(r·II _(l))  (13)

The function ℑ( ) is to remove the zero elements of a signal vector. With this operations, the length for first three segments are the same, i.e., M=2¹⁶−1. However, the last one is M′=2¹⁶−1−639×4. The demodulation process at the j-th receiver can be formulated as: D′ _(j,l) =r _(l) ·r _(j) ,l=1,2,3  (14a) D′ _(j,l) =r _(l) ·└x _(j)┘_(M′) ,l=4  (14b)

where └ ┘_(m), means the truncation of the TxID sequence to M′.

In the previous section, a simple example of the polarity modulation is used to demonstrate the principle of the proposed data transmission system. In practice, a communications system has to be designed with good spectrum efficiency, i.e. transmitting data information as much as possible. To achieve this, a higher order modulation scheme has to be used. Therefore, the signal to in-band DTV noise for the decision variable in (14) has to be analyzed before the order modulation scheme can be selected. With the similar analysis as above the signal-to-in-band DTV noise ratio can be determined for the first three decision variables as: $\begin{matrix} {{SDR}_{l} = \left\{ \begin{matrix} {{10\quad\log_{10}\frac{\rho_{l}^{2}}{E\left\{ {{\frac{1}{M}{\sum\limits_{n = 0}^{M - 1}{{d(k)}{x\left( {n + k} \right)}}}}}^{2} \right\}}},} & {{l = 1},2,3} \\ {{10\quad\log_{10}\quad\frac{\rho_{l}^{2}}{E\left\{ {{\frac{1}{M^{\prime} - 1}{\sum\limits_{n = 0}^{M^{\prime} - 1}{{d(k)}{x\left( {n + k} \right)}}}}}^{2} \right\}}},} & {l = 4} \end{matrix} \right.} & (15) \end{matrix}$

From the numerical simulations, the SDR can be determined as 18.16 and 17.99 dB for the first three decision variables and the last one. Since the difference between the SDR is very small in (15), one identical injection level can be used for the four TxID sequences within the same DTV frame to simplify the system design. From the SDR analysis in (15), it is expected 4PAM (Pulse Amplitude Modulation) can be used to increase the capacity of the proposed transmission system to 42*8 bits/s. To ensure the system performance, the bit error rate of the 4PAM is calculated according to: $\begin{matrix} {P_{4} = {\frac{6}{4}{Q\left( \sqrt{\frac{12}{15}\gamma_{b}} \right)}}} & (16) \end{matrix}$

where γ_(b) is the SNR per bit. To meet certain system performance, a desired bit error rate P₄ is first selected, e.g. 10⁻⁶ and 10⁻³ could be selected for the uncoded and coded systems, respectively. With the desired bit error rate performance, the SNR per bit γ_(b) can be determined through a table look-up approach. The correspondent two modulation levels can be calculated with the γ_(b). $\begin{matrix} {\gamma_{b} = {\frac{1}{4}{\sum\limits_{l = 1}^{4}\frac{\rho_{l,1}^{2} + \rho_{l,2}^{2}}{4\quad\sigma_{n}^{2}}}}} & (17) \end{matrix}$

with ρ_(1,2)3ρ_(l,1).

DVB-T and ISDB-T DTV system transmitters can also can be identified using a 12-bit Kasami sequence. The Kasami sequence should be locked to the FFT block for fast synchronization.

In a distributed transmission environment, if a receiving site can identify more than three transmitters, and the transmitter geographical locations as well as their DTV transmission time delays are known, the receiving location can be calculated from the differences in arrival time of the Kasami sequences. Assuming the receiver already knows the relative position of the various transmitters, as the receiver identifies the transmitter of origin of a given signal, the receiver software will be able to calculate the relative time delay between the various received signals, i.e. direct combined transmissions d_(i)′(n). From this information the receiver processor can calculate the position of the receiver relative to the transmitters.

With reference to FIG. 9, the distance between a receiver R₁ and one transmitter will establish an equation with three unknown variables of the receiver's R₁ coordinate R₁(x, y, z). Three equations are sufficient to determine the coordinate R₁(x, y, z) of the receiver R₁ when the absolute propagation time t_(i) for each transmitter is known, i.e. the location of the receiver R₁ can be determined using DTV signals from three DTV transmitters. However, when the absolute propagation time ti for each transmitter is not available, four transmitters are needed to find the coordinates of the positioning receiver R₁. Consider four transmitters, TxA to TxD, with coordinates (x_(l),y_(l),z₁) (x₂,y₂,z₂), (x₃,y₃,z₃) and (x₄,y₄,z₄), respectively, which for existing DTV transmitters are known to the positioning receiver R₁. Denoting the absolute propagation time from the i-th transmitter to the positioning receiver R₁ by t_(i), the proposed positioning algorithms can be formulated as: $\begin{matrix} \left\{ \begin{matrix} {{ct}_{1} = \sqrt{\left( {X - X_{1}} \right)^{2} + \left( {Y - Y_{1}} \right)^{2} + \left( {Z - Z_{1}} \right)^{2}}} \\ {{ct}_{2} = \sqrt{\left( {X - X_{2}} \right)^{2} + \left( {Y - Y_{2}} \right)^{2} + \left( {Z - Z_{2}} \right)^{2}}} \\ {{ct}_{3} = \sqrt{\left( {X - X_{3}} \right)^{2} + \left( {Y - Y_{3}} \right)^{2} + \left( {Z - Z_{3}} \right)^{2}}} \\ {{ct}_{4} = \sqrt{\left( {X - X_{4}} \right)^{2} + \left( {Y - Y_{4}} \right)^{2} + \left( {Z - Z_{4}} \right)^{2}}} \end{matrix} \right. & (18) \end{matrix}$

in which the constant c is the speed of light. When the absolute propagation time t_(i) for each transmitter is not available, what we know from the received signal from each transmitter is the relative propagation time t_(i)′ from each transmitter with respect to a reference time related to the transmission network. Under this circumstance, equation 18 can be re-written as: $\begin{matrix} \left\{ \begin{matrix} {{{ct}_{1}^{\prime} + {c\quad\Delta\quad t}} = \sqrt{\left( {X - X_{1}} \right)^{2} + \left( {Y - Y_{1}} \right)^{2} + \left( {Z - Z_{1}} \right)^{2}}} \\ {{{ct}_{2}^{\prime} + {c\quad\Delta\quad t}} = \sqrt{\left( {X - X_{2}} \right)^{2} + \left( {Y - Y_{2}} \right)^{2} + \left( {Z - Z_{2}} \right)^{2}}} \\ {{{ct}_{3}^{\prime} + {c\quad\Delta\quad t}} = \sqrt{\left( {X - X_{3}} \right)^{2} + \left( {Y - Y_{3}} \right)^{2} + \left( {Z - Z_{3}} \right)^{2}}} \\ {{{ct}_{4}^{\prime} + {c\quad\Delta\quad t}} = \sqrt{\left( {X - X_{4}} \right)^{2} + \left( {Y - Y_{4}} \right)^{2} + \left( {Z - Z_{4}} \right)^{2}}} \end{matrix} \right. & (19) \end{matrix}$

In which t′_(i)=t_(i)−Δt is the relative propagation time for the i-th transmitter, and Δt is the timing difference between the relative time t_(i)′ and the absolute time t_(i). The timing difference Δt is unknown, but identical to all the transmitters when all the transmitters are synchronized within the distributed transmitter network.

If the transmitters are not all synchronized, i.e. there is a timing offset between transmission times of the different transmitters, the timing offset information can be obtained via various ways, e.g. the timing offset information can transmitted using the low speed data transmission methods described above or monitor stations can be implemented to determine and broadcast the timing offsets for each transmitter. The timing offsets will depend on the individual networks characteristics, such as transmitter heights and terrain.

When there are more transmitters than needed for determining the position of the receiver, i.e. the number of the equations becomes larger than the number of the variables, the extra equations can be used to increase the positioning accuracy, and to reduce the impact of multi-path distortion, through optimization techniques. 

1. A method of identifying a transmitter in a distributed digital television transmission network, including a plurality of transmitters and a plurality of receivers, comprising the steps of: a) providing a signal to be transmitted to each transmitter; b) embedding an identification sequence into the signal, indicative of the transmitter of origin, forming a combined transmission; and c) transmitting the combined transmission from each transmitter.
 2. The method according to claim 1, further comprising: d) receiving a transmitted signal, including direct combined transmissions from each transmitter, reflected combined transmissions from each transmitter, and noise, at a reception site; and e) determining the transmitter of origin for at least one of the direct and reflected combined transmissions from the transmitted signals.
 3. The method according to claim 2, wherein the distributed digital television transmission network is a single frequency network; wherein step a) includes providing a signal to be transmitted at the same frequency to each transmitter; and wherein step e) further comprises determining the transmitter power level, and transmission time delay for each of the direct and reflected combined transmissions from the transmitted signal; and the method further comprising: f) tuning at least one of signal power level and relative time delay of at least one transmitter in accordance with results of step e) to minimize multi-path effects in intended coverage areas of the network.
 4. The method according to claim 3, further comprising repeating steps d) and e) at various locations within the network, wherein step f) is optimized for the various locations within the network.
 5. The method according to claim 2, wherein the signal is a digital television signal.
 6. The method according to claim 5, wherein step b) includes embedding the identification sequence in-band with the signal.
 7. The method according to claim 6, wherein the identification sequence is embedded at least 10 dB below a noise threshold of the transmitted signal, whereby the identification sequence has very little effect thereon.
 8. The method according to claim 5, wherein the identification sequence is time synchronized to the signal flame structure of the digital television (DTV) signal.
 9. The method according to claim 2, wherein the identification sequences are substantially-orthogonal pseudo-random sequences.
 10. The method according to claim 9, wherein step e) includes correlating the transmitted signal and a locally generated identification sequence, substantially identical to the identification sequence from one of the transmitters, to obtain a cross-correlation function, whereby when the locally generated identification sequence is identical to the identification sequence from the one transmitter, the cross-correlation function includes an impulse.
 11. The method according to claim 10, wherein step e) further comprises obtaining an impulse response for one of the transmitters from the cross-correlation function.
 12. The method according to claim 10, wherein step b) includes embedding a Kasami sequence as the identification sequence using a Kasami sequence generator.
 13. The method according to claim 12, wherein initial values of the sequence generators are similar for adjacent transmitters, thereby facilitating the determination of the transmitter of origin.
 14. The method according to claim 12, wherein step e) includes using one of: an Advanced Television Systems Committee (ATSC) DTV field, synchronization PN-511 sequence, a cyclic prefix from a Digital Video Broadcasting-Terrestrial (DVB-T), and a cyclic prefix from an Integrated Services Digital Broadcasting-Terrestrial (ISDB-T) signal for detection of the Kasami sequence and for synchronization of the Kasami sequence with the identification sequence.
 15. The method according to claim 14, wherein step e) further comprises correlating only a segment of the transmitted signal and the identification sequence from each transmitter.
 16. The method according to claim 15, wherein step e) includes a sliding window technique to select the transmitted signal for correlating with the locally generated identification sequence.
 17. The method according to claim 16, wherein the segment of the transmitted signal has a length substantially equal to a length of a DTV field plus twice a length of a delay spread of a channel impulse, whereby each segment substantially begins at a starting point of each DTV field minus one delay spread and ends at a end point of the DTV field plus one delay spread; and wherein a length of a sliding window is substantially equal to the length of one DTV field.
 18. The method according to claim 17, wherein the cross-correlation function between the segment of the transmitted signal and the identification sequence is between 40 μs and 600 μs in duration.
 19. The method according to claim 15, wherein step e) includes correlating over a plurality of segments to obtain a plurality of cross-correlation functions, and averaging the plurality of cross correlation functions to cancel out noise and improve resolution.
 20. The method according to claim 19, wherein step e) includes aligning corresponding peaks in each cross-correlation function before averaging.
 21. The method according to claim 20, wherein step e) further comprises filtering side lobes from impulses in the cross-correlation functions.
 22. The method according to claim 2, further comprising: repeating step e) to determine cross-correlation functions with impulses indicative of direct combined transmissions from at least three transmitters; determining differential time delays between the direct combined transmissions from the at least three transmitters; and determining a geographic location of the reception site based on known geographic locations of the at least three transmitters.
 23. The method according to claim 22, wherein the step of determining the geographic location of the reception site includes subtracting any offset time delays between transmitter transmission times.
 24. The method according to claim 23, wherein the identification sequences are modulated to provide information relating to the offset time delays of the transmitters.
 25. The method according to claim 22, wherein the identification sequences are substantially-orthogonal pseudo-random sequences with negligible cross-correlation.
 26. The method according to claim 25, wherein the identification sequences are selected from the group consisting of Kasami sequences and Gold sequences.
 27. The method according to claim 12, further comprising inverting the amplitude of the Kasami sequence for transmitting one bit of information per one or more Kasami sequences, whereby data is transmitted to provide information to the receiver.
 28. The method according to claim 1, wherein step e) includes embedding a plurality of identification sequences; and wherein the identification sequences are modulated to enable data to be transmitted thereby.
 30. The method according to claim 28, wherein the identification sequences are polarity modulated or pulse amplitude modulated. 